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PM6680A
Dual synchronous step-down controller with adjustable output voltages plus LDO
Features

6 V to 36 V input voltage range Adjustable output voltages 5V LDO delivers 100 mA peak current 1.237 V 1 % reference voltage available externally Current sensing using low side MOSFETs RDS(on) Valley current sensing Soft-start internally fixed at 2ms Soft output discharge Latched OVP and UVP Selectable pulse skipping at light loads Selectable minimum frequency (33 kHz) in pulse skip mode 5mW maximum quiescent power Independent power good signals Output voltage ripple compensation Thermal shutdown VFQFPN-32 5X5
Description
PM6680A is a dual step-down controller specifically designed to provide extremely high efficiency conversion, with loss less current sensing technique. The constant on-time architecture assures fast load transient response and the embedded voltage feed-forward provides nearly constant switching frequency operation. An embedded integrator control loop compensates the DC voltage error due to the output ripple. Pulse skipping technique increases efficiency at very light load. Moreover a minimum switching frequency of 33 kHz is selectable to avoid audio noise issues. The PM6680A provides a selectable switching frequency, allowing three different values of switching frequencies for the two switching sections. The output voltages OUT1 and OUT2 can be adjusted from 0.9 V to 5 V and from 0.9 V to 3.3 V respectively.
Applications

Embedded computer system FPGA system power Industrial applications on 24 V High performance and high density DC/DC modules
Table 1.
Device summary
Order codes PM6680A VFQFPN-32 5X5 (exposed pad) PM6680ATR Tape and reel Package Packaging Tube
December 2007
Rev 2
1/48
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Contents
PM6680A
Contents
1 2 Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
2.1 2.2 Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
3
Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
3.1 3.2 Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
4 5 6 7
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 Typical operating characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 Application schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 Device description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
7.1 7.2 7.3 7.4 7.5 7.6 7.7 7.8 7.9 7.10 7.11 Constant On time PWM control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Constant On time architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Output ripple compensation and loop stability . . . . . . . . . . . . . . . . . . . . . 21 Pulse skip mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 No-audible skip mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Soft start and soft end . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 Gate drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Reference voltage and bandgap . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 Internal linear regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 Power up sequencing and operative modes . . . . . . . . . . . . . . . . . . . . . . . 28
8
Monitoring and protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
2/48
PM6680A
Contents
9
Design guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
9.1 9.2 9.3 9.4 9.5 9.6 9.7 9.8 Switching frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 Output capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 Input capacitors selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32 Power MOSFETS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Closing the integrator loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 Other parts design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39 Design example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
10 11
Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
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Block diagram
PM6680A
1
Block diagram
Figure 1. Functional block diagram
VIN
VCC
REFERENCE GENERATOR
VREF
5V LINEAR REGULATOR
4V
+ -
UVLO
VREF LDO5 ENABLE
LDO5
NC
4.8V
+ UVLO
FB2 V5SW OUT2 SKIP FSEL FREQUENCY SELECTOR OUT1 BOOT2 BOOT1 FB1
HGATE2
LEVEL SHIFTER
OUT2 SMPS
OUT1 SMPS CONTROLLER
LEVEL SHIFTER
HGATE1
PHASE2
CONTROLLER
PHASE1 CSENSE1
COMP2 LDO5 LGATE2 LDO5
COMP1
LGATE1
PGOOD1
SHDN STARTUP CONTROLLER EN2 UVLO TERMIC FAULT
LDO5 ENABLE EN1 TERMIC CONTROLLER
4/48
PM6680A
Pin settings
2
2.1
Pin settings
Connections
Figure 2. Pin connection (through top view)
1
PM6680A
5/48
Pin settings
PM6680A
2.2
Functions
Table 2.
N 1 2 3 Pin SGND1 COMP2 FSEL
Pin functions
Function Signal ground. Reference for internal logic circuitry. It must be connected to the signal ground plan of the power supply. The signal ground plan and the power ground plan must be connected together in one point near the PGND pin. DC voltage error compensation pin for the switching section 2 Frequency selection pin. It provides a selectable switching frequency, allowing three different values of switching frequencies for the switching sections. Enable input for the switching section 2. * The section 2 is enabled applying a voltage greater than 2.4 V to this pin. * The section 2 is disabled applying a voltage lower than 0.8 V. When the section is disabled the High Side gate driver goes low and Low Side gate driver goes high. If both EN1 and EN2 pins are low and SHDN pin is high the device enters in standby mode. Shutdown control input. * The device switch off if the SHDN voltage is lower than the device off thershold (Shutdown mode) * The device switch on if the SHDN voltage is greater than the device on threshold. The SHDN pin can be connected to the battery through a voltage divider to program an undervoltage lockout. In shutdown mode, the gate drivers of the two switching sections are in high impedance (high-Z). Not connected. Feedback input for the switching section 2 This pin is connected to a resistive voltage-divider from OUT2 to PGND to adjust the output voltage from 0.9 V to 3.3 V. Output voltage sense for the switching section 2.This pin must be directly connected to the output votage of the switching section. Bootstrap capacitor connection for the switching section 2. It supplies the high-side gate driver. High-side gate driver ouput for section 2. This is the floating gate driver output. Switch node connection and return path for the high side driver for the section 2.It is also used as negative current sense input.
4
EN2
5
SHDN
6 7 8 9 10 11
NC FB2 OUT2 BOOT2 HGATE2 PHASE2
Positive current sense input for the switching section 2. This pin must be connected 12 CSENSE2 through a resistor to the drain of the synchronous rectifier (RDSON sensing) to obtain a positive current limit threshold for the power supply controller. 13 14 15 16 LGATE2 PGND LGATE1 SGND2 Low-side gate driver output for the section 2. Power ground. This pin must be connected to the power ground plan of the power supply. Low-side gate driver output for the section 1. Signal ground for analog circuitry. It must be connected to the signal ground plan of the power supply.
6/48
PM6680A Table 2.
N Pin
Pin settings Pin functions (continued)
Function Internal 5 V regulator bypass connection. * If V5SW is connected to OUT5 (or to an external 5 V supply) and V5SW is greater than 4.9 V, the LDO5 regulator shuts down and the LDO5 pin is directly connected to OUT5 through a 3 (max) switch. If V5SW is connected to GND, the LDO5 linear regulator is always on. 5V internal regulator output. It can provide up to 100 mA peak current. LDO5 pin supplies embedded low side gate drivers and an external load. Device supply voltage input and battery voltage sense. A bypass filter (4 and 4.7 F) between the battery and this pin is recommended.
17
V5SW
18 19
LDO5 VIN
Positive current sense input for the switching section 1. This pin must be connected 20 CSENSE1 through a resistor to the drain of the synchronous rectifier (RDSON sensing) to obtain a positive current limit threshold for the power supply controller. 21 22 23 PHASE1 HGATE1 BOOT1 Switch node connection and return path for the high side driver for the section 1.It is also used as negative current sense input. High-side gate driver ouput for section 1. This is the floating gate driver output. Bootstrap capacitor connection for the switching section 1. It supplies the high-side gate driver. Pulse skipping mode control input. * If the pin is connected to LDO5 the PWM mode is enabled. * If the pin is connected to GND, the pulse skip mode is enabled. * If the pin is connected to VREF the pulse skip mode is enabled but the switching frequency is kept higher than 33 kHz (No-audible puse skip mode). Enable input for the switching section 1. * The section 1 is enabled applying a voltage greater than 2.4 V to this pin. * The section 1 is disabled applying a voltage lower than 0.8 V. When the section is disabled the High Side gate driver goes low and Low Side gate driver goes high.
24
SKIP
25
EN1
Power Good ouput signal for the section 1. This pin is an open drain ouput and when 26 PGOOD1 the ouput of the switching section 1 is out of +/- 10 % of its nominal value.It is pulled down. Power Good ouput signal for the section 2. This pin is an open drain ouput and when 27 PGOOD2 the ouput of the switching section 2 is out of +/- 10 % of its nominal value.It is pulled down. 28 29 30 31 32 FB1 OUT1 COMP1 VCC VREF Feedback input for the switching section 1. This pin is connected to a resistive voltage-divider from OUT1 to PGND to adjust the output voltage from 0.9 V to 5.5 V. Output voltage sense for the switching section 1.This pin must be directly connected to the output votage of the switching section. DC voltage error compensation pin for the switching section 1. Device supply voltage pin. It supplies all the internal analog circuitry except the gate drivers (see LDO5). Connect this pin to LDO5. Internal 1.237 V high accuracy voltage reference. It can deliver 50 A. Bypass to SGND with a 100 nF capacitor to reduce noise.
7/48
Electrical data
PM6680A
3
3.1
Electrical data
Maximum rating
Table 3. Absolute maximum ratings
Parameter V5SW, LDO5 to PGND VIN to PGND HGATEx and BOOTx, to PHASEx PHASEx to PGND CSENSEx , to PGND CSENSEx to BOOTx LGATEx to PGND FBx, COMPx, SKIP, , FSEL,,VREF to SGND1,SGND2 PGND to SGND1,SGND2 SHDN,PGOODx, OUTx, VCC, ENx to SGND1,SGND2 Power Dissipation at TA = 25C Maximum withstanding Voltage range test condition: CDF-AEC-Q100-002- "Human Body Model" acceptance criteria: "Normal Performance"
1. PHASE to PGND up to -2.5 V for t < 10 ns 2. LGATEx to PGND up to -1 V for t < 40 ns
Value -0.3 to 6 -0.3 to 36 -0.3 to 6 -0.6
(1)
Unit V V V V V V V V V V W V
to36
-0.6 to 42 -6 to 0.3 -0.3 (2) to LDO5 +0.3 -0.3 to Vcc+0.3 -0.3 to 0.3 -0.3 to 6 2.8 VIN Other pins 1000 2000
3.2
Thermal data
Table 4.
Symbol RthJA TSTG TJ
Thermal data
Parameter Thermal resistance junction to ambient Storage temperature range Junction operating temperature range Value 35 -40 to 150 -40 to 125 Unit C/W C C
8/48
PM6680A
Electrical characteristics
4
Table 5.
Electrical characteristics
Electrical characteristics TA = -40 C to 125 C, unless otherwise specified. All parameters at operating temperature extremes are guaranteed by design and statistical analysis (not production tested).
Parameter Test condition Min Typ Max Unit
Symbol Supply section VIN VCC VV5SW
Input voltage range IC supply voltage Turn-ON voltage threshold Turn-OFF voltage threshold Hysteresis
Vout = Vref, LDO5 in regulation
5.5 4.5 4.8 4.6 20 4.75 50
36 5.5 4.9
V V V V mV
VV5SW RDS(on)
Maximum operating range LDO5 Internal bootstrap switch resistance OUTx,OUTx discharge-Mode On-resistance OUTx, OUTx discharge-Mode Synchronous rectifier Turn-on level V5SW > 4.9 V 1.8
5.5 3
V
18
25
0.2
0.36
0.6
V
Pin Ish Isb
Operating power consumption
FBx > VREF, Vref in regulation, V5WS to 5V 20 190
4 30 250
mW A A
Operating current sunk by SHDN connected to GND, VIN Operating current sunk by ENx to GND, V5SW to GND VIN
Shutdown section VSHDN Device ON threshold Device OFF threshold 1.2 0.8 1.5 0.85 1.7 0.9 V V
Soft start section Soft start ramp time Current limit and zero crossing comparator ICSENSE Input bias current limit (1) Comparator offset Zero crossing comparator offset Fixed negative current limit threshold
1. TA = -25 C to 125 C
2
3.5
ms
90 VCSENSE - VPGND VPGND - VPHASE VPGND - VPHASE -6 -1
100
110 6 11
A mV mV mV
-120
9/48
Electrical characteristics Table 5.
PM6680A
Electrical characteristics (continued) (TA = -40 C to 125 C, unless otherwise specified. All parameters at operating temperature extremes are guaranteed by design and statistical analysis (not production tested).
Parameter Test condition Min Typ Max Unit
Symbol Minimum on time
FSEL to GND On time pulse width@Vin = 24 V FSEL to VREF FSEL to LDO5 Minimum off time TOFFMIN @ Vin = 24 V Voltage reference Voltage accuracy VREF Load regulation
OUT1=3.3 V OUT2=1.8 V OUT1=3.3 V OUT2=1.8 V OUT1=3.3 V OUT2=1.8 V
595 190 400 145 300 105
700 225 470 170 355 125
805 260 545 200 410 145 ns
350
500
ns
4V < VLDO5 < 5.5 V -100 A < IREF < 100 A
1.224 -4
1.236
1.249 4 0.95
V mV mV
Undervoltage lockout fault Falling edge of REF threshold PWM comparator FB FB COMP COMP Voltage accuracy Input bias current Over voltage clamp Under voltage clamp Normal mode Pulse skip mode -909 900 0.1 250 60 -150
909
mV A mV
Line regulation Both SMPS, 6V < VIN < 36V (2) LDO5 linear regulation LDO5 linear output voltage VLDO5 LDO5 line regulation ILDO5 ULVO LDO5 current limit Under voltage lockout of LDO5 6 V < VIN < 36 V, 0 < ILDO5 < 50 mA 6 V < VIN < 36 V, ILDO5 = 20 mA , VLDO5 > UVLO 270 3.94 330 4 4.9 5.0 5.1 0.004 400 4.13 V %/V mA V 1 %
2. By demoboard test
10/48
PM6680A Table 5.
Electrical characteristics Electrical characteristics (continued) (TA = -40 C to 125 C, unless otherwise specified. All parameters at operating temperature extremes are guaranteed by design and statistical analysis (not production tested).
Parameter Test condition Min Typ Max Unit
Symbol
High and low gate drivers HGATE driver on-resistence LGATE driver on-resistance PGOOD pins UVP/OVP protections OVP UVP Over voltage threshold Under voltage threshold Upper threshold (VFB-VREF) Lower threshold (VFB-VREF) PGOOD leakage current VPGOOD1,2 forced to 5.5 V ISink = 4 mA 150 Both SMPS sections with respect to VREF 112 65 107 88 116 68 110 91 120 71 113 94 1 250 % % % % A mV HGATEx high state (pullup) HGATEx low state (pulldown) LGATEx high state (pullup) LGATEx low state (pulldown) 2.0 1.6 1.4 0.8 3 2.7 2.1 1.2
PGOOD1,2
IPGOOD1,2
VPGOOD1,2 Output low voltage
Thermal shutdown TSDN Shutdown temperature 150 C
Power management pins EN1,2 SMPS disabled level SMPS enabled level Low level (3) FSEL Frequency selection range Middle level (3) High level (3) Pulse skip mode SKIP PWM mode Ultrasonic mode
(3) (3)
0.8 2.4 0.5 1.0 VLDO50.8 0.5 1.0 VLDO50.8 1 1 1 1 VLDO51.5 VLDO51.5
V
V
V
(3)
VEN1,2 = 0 to 5 V Input leakage current VSKIP = 0 to 5 V VSHDN = 0 to 5 V VFSEL = 0 to 5 V
3. By design
A
11/48
Typical operating characteristics
PM6680A
5
Typical operating characteristics
FSEL=GND(200/300 kHz), SKIP=GND(skip mode), V5SW=EXT5V (external 5 V power supply connected), input voltage VIN = 24 V, SHDN, EN1 and EN2 high, OUT1 = 3.3 V, OUT2 = 1.8 V, no load unless specified)
Figure 3.
OUT1 = 3.3 V efficiency
Figure 4.
OUT2 = 1.8 V efficiency
Figure 5.
PWM no load battery current vs input voltage
Figure 6.
Skip no load battery current vs input voltage
12/48
PM6680A
Typical operating characteristics
Figure 7.
No-audible skip no load battery current vs input voltage
Figure 8.
Standby mode input battery current vs input voltage
Figure 9.
Shutdown mode input battery current vs input voltage
Figure 10. LDO5 vs output current
Figure 11. OUT1 = 3.3 V switching frequency
Figure 12. OUT2 = 1.8 V switching frequency
13/48
Typical operating characteristics
PM6680A
Figure 13. OUT1 = 3.3 V load regulation
Figure 14. OUT2 = 1.8 V load regulation
Figure 15. Voltage reference vs load current
Figure 16. OUT1, OUT2 and LDO5 Power-Up
Figure 17. OUT1 = 3.3V load transient 02A
Figure 18. OUT2 = 1.8V load transient 02A
14/48
PM6680A
Typical operating characteristics
Figure 19. 3.3 V soft start (1 load)
Figure 20. 1.8 V soft start (0.6 load)
Figure 21. OUT1 = 3.3 V soft end (no load)
Figure 22. OUT2 = 1.8 V soft end (no load)
Figure 23. OUT1 = 3.3 V soft end (0.8 load)
Figure 24. OUT2 = 1.8 V soft end (0.6 load)
15/48
Typical operating characteristics
PM6680A
Figure 25. 3.3 V no-audible skip mode
Figure 26. 1.8 V no-audible skip mode
16/48
6
PM6680A
J3 LDO5V+ VIN J5 D1 V+ BOOT1 R26 +VIN J2 -VIN PGND R5 R11 BOOT2 R3 5 6 7 8 C5 M1 4 L1 1 2 3 1 R20 22 HGATE1 PM6680A PHASE2 M3 15 LGATE1 1 2 3 CSENSE1 V5SW OUT1 COMP1 SGND PGOOD1 PGOOD2 SHDN FB2 7 S13 2 FB1 28 FB1 1 NC 6 1 S12 2 COMP2 2 C17 R30 SGND OUT2 8 C15 SGND 1 SGND PGND R25 PGND 14 PGND CSENSE2 12 R7 D3 C9 C10 V5SW OUT1 30 LDO_FB 16 26 27 5 C16 V+ V+ J6 PGOOD1 J7 EN2 EN1 VREF SKIP FSEL PGOOD2 4 25 32 24 3 C27 1uF S1 SGND LDO_FB V+ C23 V+ R16 R14 V+ 8 7 6 5 S2 SGND 1 2 3 4 SGND 2 3 4 2 3 S10 CREF 1 SGND 4 S3 SGND R15 2 1 3 SGND R31 R13 R12 29 17 LGATE2 R21 4 R8 20 13 11 5 6 7 8 R18 C24 PHASE1 HGATE2 21 R23 4 3 2 1 10 C13 2 S4 J1
+ +
R9 C20 C21 RLD5V C19 C26 C1 C2
+ CIN
VIN V+ BOOT2 C3 C22 SGND PGND 31 18 R4 8 7 6 5 VCC LDO5 VIN C6 R22 4 3 2 1 BOOT1 BOOT2 23 9 19 BOOT1 U1 R10 PGND V+ SGND SGND SGND C4 SGND SGND PGND PGND PGND
R6
C14
OUT1+ M2
S5
J4
2
L2
1
8 7 6 5
1
1
3
OUT1+ R19 D2
M4
3 S8
OUT2+
S9
+C7
+C8
2
Application schematic
C25
+ C11
+ C12
2
J9
J8
OUT2-
Figure 27. Simplified application schematic
OUT1-
PGND
R24
J10
C18 VIN
PGND
R28 SGND R17
PGND
+
C28 R33 R32 SGND SGND SGND LDO_ADJ R29
FB1
R27
SGND
V5SW OUT1+ 2 3 4 EXT5V
J11
S11
SGND 1
SGND
SGND
Application schematic
SGND
1
17/48
Device description
PM6680A
7
Device description
The PM6680A is a dual step-down controller dedicated to provide logic voltages for industrial automation applications. It is based on a Constant On Time control architecture. This type of control offers a very fast load transient response with a minimum external component count. A typical application circuit is shown in Figure 3. The PM6680A regulates two adjustable output voltages: OUT1 and OUT2. The switching frequency of the two sections can be adjusted to 200/300 kHz, 300/400 kHz or 400/500 kHz respectively. In order to maximize the efficiency at light load condition, a pulse skipping mode can be selected. The PM6680A includes also a 5 V linear regulator (LDO5) that can power the switching drivers. If the output OUT1 regulates 5 V, in order to maximize the efficiency in higher consumption status, the linear regulator can be turned off and their outputs can be supplied directly from the switching outputs. The PM6680A provides protection versus overvoltage, undervoltage and over temperature as well as power good signals for monitoring purposes. An external 1.237 V reference is available.
7.1
Constant on time PWM control
If the SKIP pin is tied to 5 V, the device works in PWM mode. Each power section has an independent on time control.The PM6680A employees a pseudo-fixed switching frequency, Constant On Time (COT) controller as core of the switched mode section. Each power section has an independent COT control. The COT controller is based on a relatively simple algorithm and uses the ripple voltage due to the output capacitor's ESR to trigger the fixed on-time one-shot generator. In this way, the output capacitor's ESR acts as a current sense resistor providing the appropriate ramp signal to the PWM comparator. On-time one-shot duration is directly proportional to the output voltage, sensed at the OUT1/OUT2 pins, and inversely proportional to the input voltage, sensed at the VIN pin, as follows: Equation 1
V OUT T ON = K -------------V IN
This leads to a nearly constant switching frequency, regardless of input and output voltages. When the output voltage goes lower than the regulated voltage Vreg, the on-time one shot generator directly drives the high side MOSFET for a fixed on time allowing the inductor current to increase; after the on time, an off time phase, in which the low side MOSFET is turned on, follows. Figure 28 shows the inductor current and the output voltage waveforms in PWM mode.
18/48
PM6680A Figure 28. Constant ON time PWM control
Device description
The duty cycle of the buck converter in steady state is: Equation 2
V OUT D = -------------V IN
The PWM control works at a nearly fixed frequency fSW: Equation 3
V OUT -------------V IN ------------------------------ = 1 K on f SW = V OUT K on x -------------V IN
As mentioned the steady state switching frequency is theoretically independent from input voltage and from output voltage. Actually the frequency depends on parasitic voltage drops that are present during the charging path(high side switch resistance, inductor resistance(DCR)) and discharging path(low side switch resistance, DCR). As a result the switching frequency increases as a function of the load current. Standard switching frequency values can be selected for both sections by pin FSEL as shown in the following table: Table 6. FSEL pin selection: typical switching frequency
Fsw@OUT1 = 3.3 V (kHz) FSEL = GND FSEL = VREF FSEL = LDO5 195 295 390 Fsw@OUT2 = 1.8 V (kHz) 335 440 600
19/48
Device description
PM6680A
7.2
Constant on time architecture
Figure 29 shows the simplified block diagram of a constant on time controller. A minimum off-time constrain (350 ns typ.) is introduced to allow inductor valley current sensing on synchronous switch. A minimum on-time (130 ns) is also introduced to assure the start-up switching sequence. PM6680A has a one-shot generator for each power section that turns on the high side MOSFET when the following conditions are satisfied simultaneously: the PWM comparator is high, the synchronous rectifier current is below the current limit threshold, and the minimum off-time has timed out. Once the on-time has timed out, the high side switch is turned off, while the synchronous switch is turned on according to the anti-cross conduction circuitry management. When the negative input voltage at the PWM comparator (Figure 29), which is a scaleddown replica of the output voltage (see the external R1/R2 divider in Figure 29), reaches the valley limit (determined by internal reference Vr = 0.9 V), the low-side MOSFET is turned off according to the anti-cross conduction logic once again, and a new cycle begins. Figure 29. Constant on-time block diagram
In steady state the FB pin voltage is about Vr and the regulated output voltage depends on the external divider: Equation 4
R2 OUT = Vr x 1 + ------ R
1
20/48
PM6680A
Device description
7.3
Output ripple compensation and loop stability
In a classic constant on time control, the system regulates the valley value of the output voltage and not the average value, as shown in Figure 28 In this condition, the output voltage ripple is source of a DC static error. To compensate this error, an integrator network can be introduced in the control loop, by connecting the output voltage to the COMP1/COMP2 (for the OUT1 and OUT2 sections respectively) pin through a capacitor CINT as in Figure 30. Figure 30. Circuitry for output ripple compensation
The integrator amplifier generates a current, proportional to the DC errors between the FB voltage and Vr, which decreases the output voltage in order to compensate the total static error, including the voltage drop on PCB traces. In addition, CINT provides an AC path for the output ripple. In steady state, the voltage on COMP1/COMP2 pin is the sum of the reference voltage Vr and the output ripple (see Figure 30). In fact when the voltage on the COMP pin reaches Vr, a fixed Ton begins and the output increases. For example, we consider Vout = 5 V with an output ripple of V = 50 mV. Considering CINT >> CFILT, the CINT DC voltage drop VCINT is about 5 V -Vr + 25 mV = 4.125 V. CINT assures an AC path for the output voltage ripple. Then the COMP pin ripple is a replica of the output ripple, with a DC value of Vr + 25 mV = 925 mV. For more details about the output ripple compensation network, see the Chapter 9.6: Closing the integrator loop on page 35 in the Design guidelines.
21/48
Device description
PM6680A
7.4
Pulse skip mode
If the SKIP pin is tied to ground, the device works in skip mode. At light loads a zero-crossing comparator truncates the low-side switch on-time when the inductor current becomes negative. In this condition the section works in discontinuous conduction mode. The threshold between continuous and discontinuous conduction mode is: Equation 5
V IN - V OUT ILOAD ( SKIP ) = ----------------------------- x T ON 2xL
For higher loads the inductor current doesn't cross the zero and the device works in the same way as in PWM mode and the frequency is fixed to the nominal value. Figure 31. PWM and pulse skip mode inductor current
Figure 31 shows inductor current waveforms in PWM and SKIP mode. In order to keep average inductor current equal to load current, in SKIP mode some switching cycles are skipped. When the output ripple reaches the regulated voltage Vreg, a new cycle begins. The off cycle duration and the switching frequency depend on the load condition. As a result of the control technique, losses are reduced at light loads, improving the system efficiency.
22/48
PM6680A
Device description
7.5
No-audible skip mode
If SKIP pin is tied to VREF, a no-audible skip mode with a minimum switching frequency of 33 kHz is enabled. At light load condition, If there is not a new switching cycle within a 30 s (typ.) period, a no-audible skip mode cycle begins. Figure 32. No audible skip mode
The low side switch is turned on until the output voltage crosses about Vreg + 1 %. Then the high side MOSFET is turned on for a fixed on time period. Afterwards the low side switch is enabled until the inductor current reaches the zero-crossing threshold. This keeps the switching frequency higher than 33 kHz. As a consequence of the control, the regulated voltage can be slightly higher than Vreg (up to 1 % ). If, due to the load, the frequency is higher than 33 kHz, the device works like in skip mode. No-audible skip mode reduces audio frequency noise that may occur in pulse skip mode at very light loads, keeping the efficiency higher than in PWM mode.
23/48
Device description
PM6680A
7.6
Current limit
The current-limit circuit employs a "valley" current-sensing algorithm. During the conduction time of the low side MOSFET the current flowing through it is sensed. The current-sensing element is the low side MOSFET on-resistance (Figure 33). Figure 33. Rsense sensing technique
HS
HGATE
PHASE
Rcsense
CSENSE
LGATE
LS RDSon
An internal 100 A current source is connected to CSENSE pin and determines a voltage drop on RCSENSE. If the voltage across the sensing element is greater than this voltage drop, the controller doesn't initiate a new cycle. A new cycle starts only when the sensed current goes below the current limit. Since the current limit circuit is a valley current limit, the actual peak current limit is greater than the current limit threshold by an amount equal to the inductor ripple current. Moreover the maximum DC load is equal to the valley current limit plus half of the inductor ripple current: Equation 6
ILOAD (max) = ILvalley +
IL 2
The output current limit depends on the current ripple, as shown in Figure 34: Figure 34. Current waveforms in current limit conditions
24/48
PM6680A
Device description
Being fixed the valley threshold, the greater the current ripple is, greater the DC output current is The valley current limit can be set with resistor RCSENSE:
Equation 7
R DS ( on ) x I Lvalley R CSENSE = --------------------------------------------Icsense
Where ICSENSE = 100 A, RDSon is the drain-source on resistance of the low side switch. Consider the temperature effect and the worst case value in RDSon calculation. The accuracy of the valley current threshold detection depends on the offset of the internal comparator (VOFF) and on the accuracy of the current generator (ICSENSE)
Equation 8
I Lvalley I Lvalley
=
RCSENSE R SNS I CSENSE VOFF + x 100 + + I CSENSE R SNS RCSENSE x I CSENSE RCSENSE
Where RSNS is the sensing element(RDSon) PM6680A provides also a fixed negative peak current limit to prevent an excessive reverse inductor current when the switching section sinks current from the load in PWM mode. This negative current limit threshold is measured between PHASE and SGND pins, comparing the magnitude drop on the PHASE node during the conduction time of the low side MOSFET with an internal fixed voltage of 120 mV. The negative valley-current limit INEG (if the device works in PWM mode) is given by:
Equation 9
I NEG =
120mV RDSon
7.7
Soft start and soft end
Each switching section is enabled separately by asserting high EN1/EN2 pins respectively. In order to realize the soft start, at the startup the overcurrent threshold is set 25 % of the nominal value and the undervoltage protection (see related sections) is disabled. The controller starts charging the output capacitor working in current limit. The overcurrent threshold is increased from 25 % to 100 % of the nominal value with steps of 25 % every 700 s (typ.). After 2.8 ms (typ.) the undervoltage protection is enabled. The soft start time is not programmable. A minimum capacitor CINT is required to ensure a soft start without any overshoot on the output:
Equation 10
CINT 6A x C out ILvalley IL + 4 2
25/48
Device description Figure 35. Soft start waveforms
PM6680A
When a switching section is turned off (EN1/EN2 pins low), the controller enters in soft end mode.The output capacitor is discharged through an internal 18 p-MOSFET switch; when the output voltage reaches 0.3 V, the low-side MOSFET turns on, keeping the output to ground. The soft end time also depends on load condition.
7.8
Gate drivers
The integrated high-current drivers allow to use different power MOSFETs. The high side driver MOSFET uses a bootstrap circuit which is indirectly supplied by LDO5 output. The BOOT and PHASE pins work respectively as supply and return rails for the HS driver. The low side driver uses the internal LDO5 output for the supply rail and PGND pin as return rail. An important feature of the gate drivers is the adaptive anti-cross conduction protection, which prevents high side and low side MOSFETs from being on at the same time. When the high side MOSFET is turned off the voltage at the phase node begins to fall. The low side MOSFET is turned on when the voltage at the phase node reaches an internal threshold. When the low side MOSFET is turned off, the high side remains off until the LGATE pin voltage goes approximatively under 1 V. The power dissipation of the drivers is a function of the total gate charge of the external power MOSFETs and the switching frequency, as shown in the following equation:
Equation 11
P driver = V driver x Q g x f SW
Where Vdriver is the 5 V driver supply.
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PM6680A
Device description
7.9
Reference voltage and bandgap
The 1.237 V (typ.) internal bandgap voltage is accurate to 1 % over the temperature range. It is externally available (VREF pin) and can supply up to 100 A and can be used as a voltage threshold for the multifunction pins FSEL and SKIP to select the appropriate working mode. Bypass VREF to ground with a 100 nF minimum capacitor. If VREF goes below 0.87 V (typ.) , the system detects a fault condition and all the circuitry is turned off. A toggle on the input voltage (power on reset) or a toggle on SHDN pin is necessary to restart the device. An internal divider of the bandgap provides a voltage reference Vr of 0.9 V. This voltage is used as reference for the linear and the switching regulators outputs. The overvoltage protection, the undervoltage protection and the power good signals are referred to Vr.
7.10
Internal linear regulator
The PM6680A has an internal linear regulator providing 5 V (LDO5) at 2 % accuracy. High side drivers, low side drivers and most of internal circuitry are supplied by LDO5 output through VCC pin (an external RC filter may be applied between LDO5 and VCC). The linear regulator can provide an average output current of 50 mA and a peak output current of 100 mA. Bypass LDO5 output with a minimum 1F ceramic capacitor and a 4,7 F tantalum capacitor ( ESR 2 ). If the 5 V output goes below 4 V, the system detects a fault condition and all the circuitry is turned off. A power on reset or a toggle on SHDN pin is necessary to restart the device. V5SW pin allows to keep the 5 V linear regulator always active or to enable the internal bootstrap-switchover function: if the 5 V switching output is connected to V5SW, when the voltage on V5SW pin is above 4.8 V, an internal 3.0 max p-channel MOSFET switch connects V5SW pin to LDO5 pin and simultaneously LDO5 shuts down. This configuration allows to achieve higher efficiency. V5SW can be connected also to an external 5 V supply. LDO5 regulator turns off and LDO5 is supplied externally. If V5SW is connected to ground, the internal 5 V regulator is always on and supplies LDO5 output Table 7.
V5SW GND
V5SW multifunction pin
Description The 5 V linear regulator is always turned on and supplies LDO5 output.
Switching 5 V The 5 V linear regulator is turned off when the voltage on V5SW is above 4.8 V and output the LDO5 output is supplied by the switching 5 V output. External 5 V supply The 5 V linear regulator is turned off when the voltage on V5SW is above 4.8 V and LDO5 output is supplied by the external 5 V.
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Device description
PM6680A
7.11
Power up sequencing and operative modes
Let us consider SHDN, EN1 and EN2 low at the beginning. An external voltage is applied as input voltage. The device is in shutdown mode. When the SHDN pin voltage is above the shutdown device on threshold (1.5 V typ.), the controller begins the power-up sequence. All the latched faults are cleared. LDO5 undervoltage control is blanked for 4 ms and the internal regulator LDO5 turns on. If the LDO5 output is above the UVLO threshold after this time, the device enters in standby mode. The switching outputs are kept to ground by turning on the low side MOSFETs. When EN1 and EN2 pins are forced high the switching sections begin their soft start sequence.
Table 8.
Mode Run
Operatives modes
Conditions SHDN is high, EN1/EN2 pins are high Description Switching regulators are enabled; internal linear regulators outputs are enabled.
Standby Shutdown
Internal Linear regulators active (LDO5 is always on). Both EN1/EN2 pins are low In Standby mode LGATE1/LGATE2 pins are forced and SHDN pin is high high while HGATE1/HGATE2 pins are forced low. SHDN is low All circuits off.
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PM6680A
Monitoring and protections
8
Monitoring and protections
Power good signals
The PM6680A provides two independent power good signals: one for each switching section (PGOOD1/PGOOD2). PGOOD1/PGOOD2 signals are low if the output voltage is out of 10 % of the designed set point or during the soft-start, standby and shutdown mode.
Thermal protection
The PM6680A has a thermal protection to preserve the device from overheating. The thermal shutdown occurs when the die temperature goes above +150 C. In this case all internal circutry is turned off and the power sections are turned off after the discharge mode. A power on reset or a toggle on the SHDN pin is necessary to restart the device.
Overvoltage protection
When the switching output voltage is about 115 % of its nominal value, a latched overvoltage protection occurs. In this case, the synchronous rectifier immediately turns on while the high-side MOSFET turns off. The output capacitor is rapidly discharged and the load is preserved from being damaged. The overvoltge protection is also active during the soft start. Once an overvoltage protection has been detected, a toggle on SHDN, EN1/EN2 pins or a power on reset is necessary to exit from the latched state.
Undervoltage protection
When the switching output voltage is below 70 % of its nominal value, a latched undervoltage protection occurs. In this case the switching section is immediately disabled and both switches are open. The controller enters in soft end mode and the output is eventually kept to ground, turning low side MOSFET on. The undervoltage circuit protection is enabled only at the end of the soft-start. Once an overvoltage protection has been detected, a toggle on SHDN, EN1/EN2 pin or a power on reset is necessary to clear the undervoltage fault and starts with a new soft-start phase.
Table 9.
Mode Overvoltage protection Undervoltage protection Thermal shutdown
Protections and operatives modes
Conditions Description
LGATE1/LGATE2 pin is forced high, LDO5 remains OUT1/OUT2 > 115% of the active. Exit by a power on reset or toggling SHDN or nominal value EN1/EN2 LGATE1/LGATE2 is forced high after the soft end OUT1/OUT2 < 70 % of the mode, LDO5 remains active. Exit by a power on reset nominal value or toggling SHDN or EN1/EN2 TJ> +150 C All circuitry off. Exit by a POR on VIN or toggling SHDN.
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Design guidelines
PM6680A
9
Design guidelines
The design of a switching section starts from two parameters:

Input voltage range: in notebook applications it varies from the minimum battery voltage, VINmin to the AC adapter voltage, VINmax. Maximum load current: it is the maximum required output current, ILOAD(max).
9.1
Switching frequency
It's possible to set 3 different working frequency ranges for the two sections with FSEL pin (Table 6). Switching frequency mainly influences two parameters:

Inductor size: for a given saturation current and RMS current, greater frequency allows to use lower inductor values, which means smaller size. Efficiency: switching losses are proportional to frequency. High frequency generally involves low efficiency.
9.2
Inductor selection
Once that switching frequency is defined, inductor selection depends on the desired inductor ripple current and load transient performance. Low inductance means great ripple current and could generate great output noise. On the other hand, low inductor values involve fast load transient response. A good compromise between the transient response time, the efficiency, the cost and the size is to choose the inductor value in order to maintain the inductor ripple current IL between 20 % and 50 % of the maximum output current ILOAD(max). The maximum IL occurs at the maximum input voltage. With this considerations, the inductor value can be calculated with the following relationship:
Equation 12
L=
VIN - VOUT VOUT x fsw x IL VIN
where fsw is the switching frequency, VIN is the input voltage, VOUT is the output voltage and IL is the selected inductor ripple current. In order to prevent overtemperature working conditions, inductor must be able to provide an RMS current greater than the maximum RMS inductor current ILRMS:
Equation 13
ILRMS = (ILOAD (max)) 2 + Where IL(max) is the maximum ripple current:
30/48
(IL (max)) 2 12
PM6680A Equation 14
Design guidelines
IL (max) =
VINmax - VOUT VOUT x fsw x L VINmax
If hard saturation inductors are used, the inductor saturation current should be much greater than the maximum inductor peak current Ipeak:
Equation 15
Ipeak = ILOAD (max) +
IL (max) 2
Using soft saturation inductors it's possible to choose inductors with saturation current limit nearly to Ipeak. Below there is a list of some inductor manufacturers.
Table 10. Inductor manufacturer
Series MSS1038 MSS7341 TPC Inductor value (uH) 1.5 to 22 3.3 to 22 1 to 22 H RMS current (A) 2.85 to 7.85 1.7 to 3.95 2.7 to 8 Saturation current (A) 2.9 to 8.30 1.3 to 3.5 2.6 to 9.5
Manufacturer COILCRAFT COILCRAFT WURTH
9.3
Output capacitor
The selection of the output capacitor is based on the ESR value Rout and the voltage rating rather than on the capacitor value Cout. The output capacitor has to satisfy the output voltage ripple requirements. Lower inductor value can reduce the size of the choke but increases the inductor current ripple IL. Since the voltage ripple VRIPPLEout is given by:
Equation 16
VRIPPLEout = R out x IL
A low ESR capacitor is required to reduce the output voltage ripple. Switching sections can work correctly even with 20 mV output ripple. However, to reduce jitter noise between the two switching sections it's preferable to work with an output voltage ripple greater than 30 mV. If lower output ripple is required, a further compensation network is needed (see Closing the integrator loop paragraph). Finally the output capacitor choice deeply impacts on the load transient response (see Load transient response paragraph). Below there is a list of some capacitor manufacturers.
31/48
Design guidelines
PM6680A
Table 11.
Output capacitor manufacturer
Series POSCAP TPB, TPD, TPE SPCAP UD, UE Capacitor value (uF) 100 to 470 100 to 470 Rated voltage (V) 2.5 to 6.3 2 to 6.3 ESR max (m) 12 to 65 7 to 18
Manufacturer SANYO PANASONIC
9.4
Input capacitors selection
In a buck topology converter the current that flows into the input capacitor is a pulsed current with zero average value. The input RMS current of the two switching sections can be roughly estimated as follows:
Equation 17
2 ICinRMS = D1 x I1 x (1 - D1) + D 2 x I2 x (1 - D 2 ) 2
Where D1, D2 are the duty cycles and I1, I2 are the maximum load currents of the two sections. Input capacitor should be chosen with an RMS rated current higher than the maximum RMS current given by both sections. Tantalum capacitors are good in term of low ESR and small size, but they occasionally can burn out if subjected to very high current during the charge. Ceramic capacitors have usually a higher RMS current rating with smaller size and they remain the best choice. Below there is a list of some ceramic capacitor manufacturers.
Table 12. Input capacitor manufacturer
Series UMK325BJ106KM-T GMK325BJ106MN Capacitor value (uF) 10 10 Rated voltage (V) 50 35
Manufacturer TAYIO YUDEN TAYIO YUDEN
32/48
PM6680A
Design guidelines
9.5
Power MOSFETS
Logic-level MOSFETs are recommended, since low side and high side gate drivers are powered by LDO5. Their breakdown voltage VBRDSS must be higher than VINmax. In notebook applications, power management efficiency is a high level requirement. The power dissipation on the power switches becomes an important factor in switching selections. Losses of high-side and low-side MOSFETs depend on their working conditions. The power dissipation of the high-side MOSFET is given by:
Equation 18
PDHighSide = Pconduction + Pswitching
Maximum conduction losses are approximately:
Equation 19
Pconduction = RDSon x
VOUT x ILOAD (max)2 VINmin
where RDSon is the drain-source on resistance of the high side MOSFET. Switching losses are approximately:
Equation 20
Pswitching =
VIN x (ILOAD (max) -
IL I ) x t on x fsw VIN x (ILOAD (max) + L ) x t off x fsw 2 2 + 2 2
where ton and toff are the switching times of the turn on and turn off phases of the MOSFET. As general rule, high side MOSFETs with low gate charge are recommended, in order to minimize driver losses. Below there is a list of possible choices for the high side MOSFET.
Table 13. High side MOSFET manufacturer
Type STS5NF60L Gate charge (nC) 25 Rated reverse voltage (V) 60
Manufacturer ST
The power dissipation of the low side MOSFET is given by:
Equation 21
PDLowSide = Pconduction
Maximum conduction losses occur at the maximum input voltage:
33/48
Design guidelines Equation 22
PM6680A
V Pconduction = RDSon x 1 - OUT V IN max
x ILOAD (max) 2
Choose a synchronous rectifier with low RDSon. When high side MOSFET turns on, the fast variation of the phase node voltage can bring up even the low side gate through its gatedrain capacitance CRSS, causing cross-conduction problems. Choose a low side MOSFET that minimizes the ratio CRSS/CGS (CGS = CISS - CRSS). Below there is a list of some possible low side MOSFETs.
Table 10. Low side MOSFET manufacturer
Manufacturer ST Type STS7NF60L RDSon (m) [VC11] 19 C RSS -------------C GS 0.0625 Rated reverse voltage (V) 60
Dual n-channel MOSFETs can be used in applications with a maximum output current of about 3 A. Below there is a list of some MOSFET manufacturers.
Table 14. Dual MOSFET manufacturer
Type STS4DNF60L RDSon (m) 50 Gate charge (nC) 15 Rated reverse voltage (V) 60
Manufacturer ST
A rectifier across the low side MOSFET is recommended. The rectifier works as a voltage clamp across the synchronous rectifier and reduces the negative inductor swing during the dead time between turning the high-side MOSFET off and the synchronous rectifier on. It can increase the efficiency of the switching section, since it reduces the low side switch losses. A shottky diode is suitable for its low forward voltage drop (0.3 V). The diode reverse voltage must be greater than the maximum input voltage VINmax. A minimum recovery reverse charge is preferable. Below there is a list of some shottky diode manufacturers.
Table 15. Schottky diode manufacturer
Series STPS1L40M Forward voltage (V) 0.5 Rated reverse voltage (V) 40 Reverse current (uA) 21
Manufacturer ST
34/48
PM6680A
Design guidelines
9.6
Closing the integrator loop
The design of external feedback network depends on the output voltage ripple. If the ripple is higher than approximately 30 mV, the feedback network (Figure 36) is usually enough to keep the loop stable.
Figure 36. Circuitry for output ripple compensation
COMP PIN VOLTAGE
Vr t
OUTPUT VOLTAGE
V ?
COMP CFILT CINT RINT
I=gm(V1-Vr)
Vr
+ -
PWM Comparator
V ?
gm
VCINT
Vr
+
V1
t
L ROUT COUT
OUT R2 FB R1
D
The stability of the system depends firstly on the output capacitor zero frequency. The following condition should be satisfied:
Equation 23
fsw > k x fZout =
k 2 x C out x R out
where k is a design parameter greater than 3 and Rout is the ESR of the output capacitor. It determinates the minimum integrator capacitor value CINT:
Equation 24
CINT >
gm f 2 x sw - fZout k
x
Vr VOUT
where gm = 50 s is the integrator transconductance.
35/48
Design guidelines
PM6680A
In order to ensure stability it must be also verified that:
Equation 25
CINT >
gm Vr x 2 x fZout VOUT
In order to reduce ground noise due to load transient on the other section, it is recommended to add a resistor RINT and a capacitor Cfilt that, together with CINT, realize a low pass filter (see figure 13). The cutoff frequency fCUT must be much greater (10 or more times) than the switching frequency of the section:
Equation 26
RINT =
1 C x C filt 2 x fCUT x INT CINT + C filt
Due to the capacitive divider (CINT, Cfilt), the ripple voltage at the COMP pin is given by:
Equation 27
VRIPPLEINT = VRIPPLEout x
CINT = VRIPPLEout x q CINT + C filt
Where VRIPPLEout is the output ripple and q is the attenuation factor of the output ripple. If the ripple is very small (lower than approximately 30 mV), a further compensation network, named virtual ESR network, is needed. This additional part generates a triangular ripple that is added to the ESR output voltage ripple at the input of the integrator network. The complete control schematic is represented in Figure 37.
36/48
PM6680A Figure 37. Virtual ESR network
T node voltage V1 Output voltage V t COMP pin voltage Vr t V1
Design guidelines
CFILT
Vr
COMP
+ - PWM
t
R1
T
RINT
CINT
gm
Comparator
+
V1
Vr
-
C L
R R2 R1
OU T ROUT FB
D
COUT
The T node voltage is the sum of the output voltage and the triangular waveform generated by the virtual ESR network. In fact the virtual ESR network behaves like a further equivalent ESR. A good trade-off is to design the network in order to achieve an RESR given by:
Equation 28
RESR =
VRIPPLE - R out IL
where IL is the inductor current ripple and VRIPPLE is the overall ripple of the T node voltage. It should be chosen higher than approximately 30 mV. The new closed loop gain depends on CINT. In order to ensure stability it must be verified that:
Equation 29
CINT >
gm Vr x 2 x fZ VOUT
Where:
37/48
Design guidelines Equation 30
PM6680A
fZ =
1 2 x C out x R TOT
where RTOT is the sum of the ESR of the output capacitor Rout and the equivalent ESR given by the virtual ESR network RESR. Moreover CINT must meet the following condition:
Equation 31
fsw > k x fZ =
k 2 x C out x R TOT
Where k is a free design parameter greater than 3 and determines the minimum integrator capacitor value CINT:
Equation 32
CINT >
gm Vr x fsw VOUT 2 x k - fZ
C must be selected as shown:
Equation 33
C > 5 x CINT
R must be chosen in order to have enough ripple voltage on integrator input:
Equation 34
R=
R1 can be selected as follows:
Equation 35
L RESR x C
1 Rx Cxx f Z R1 = 1 R- C x x fZ
Example: OUT1=1.5 V, fSW = 290 kHz, L = 2.5 H, Cout = 330 F with Rout < 12 m. We design RESR = 12 m. We choose CINT = 1 nF by equations 30, 33 and Cfilt = 47 pF, RINT = 1 k by eq.27, 28. C = 5.6 nF by Eq.34. Then R = 36 k (eq.34) and R1 = 3 k (eq.35).
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PM6680A
Design guidelines
9.7
Other parts design
VIN filter A VIN pin low pass filter is suggested to reduce switching noise. The low pass filter is shown in the next figure:
Figure 38. VIN pin filter
R
Input voltage
VIN C 100pF
Typical components values are: R = 3.9 and C = 4.7 F.
VCC filter A VCC low pass filter helps to reject switching commutations noise:
Figure 39. Inductor current waveforms
LDO5 R C VCC
Typical components values are: R = 47 and C = 1F.
VREF capacitor A 10nF to 100nF ceramic capacitor on VREF pin must be added to ensure noise rejection.
LDO5 output capacitors Bypass the output of each linear regulator with 1 F ceramic capacitor closer to the LDO pin and a 4.7F tantalum capacitor (ESR = 2 ). In most applicative conditions a 4.7 F ceramic output capacitor can be enough to ensure stability.
Bootstrap circuit The external bootstrap circuit is represented in the next figure:
39/48
Design guidelines Figure 40. Bootstrap circuit
D RBOOT CBOOT LDO5 BOOT PHASE
PM6680A
L
The bootstrap circuit capacitor value CBOOT must provide the total gate charge to the high side MOSFET during turn on phase. A typical value is 100 nF. The bootstrap diode D must charge the capacitor during the off time phases. The maximum rated voltage must be higher than VINmax. A resistor RBOOT on the BOOT pin could be added in order to reduce noise when the phase node rises up, working like a gate resistor for the turn on phase of the high side MOSFET.
9.8
Design example
The following design example considers an input voltage from 16 V to 32 V(the typical value is 24 V). The two switching outputs are OUT1 = 3.3 V and OUT2 = 1.8 V and must deliver a maximum current of 2.5 A. The selected switching frequencies are about 290 kHz for OUT1 section and about 440 kHz for OUT2 section (see Table 6).
1. Inductor selection
OUT1: ILOAD = 2.5 A, 45 % ripple current.
Equation 36
We choose standard value L= 8.2 H. IL(max) = 1.16 A @VIN = 24 V. ILRMS = 2.523 A Ipeak = 2.5 A + 0.58 A = 3.83 A OUT2:ILOAD=2.5 A, 35 % ripple current.
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PM6680A Equation 37
Design guidelines
We choose standard value L=4.7 H. IL(max) = 0.886 A @VIN =24 V. ILRMS = 2.523 A Ipeak = 2.5 A + 0.58 A = 3.83 A
2. Output capacitor selection
We would like to have an output ripple smaller than 25 mV. OUT1: POSCAP 4TPE150MI OUT2: POSCAP 6TPE220M
3. Power MOSFETs
OUT1:High side: STS5NF60L Low side: STS7NF60L OUT2:High side: STS5NF60L Low side: STS7NF60L
4. Current limit
OUT1:
Equation 38
A
Equation 39
(Let's assume the maximum temperature Tmax = 75 C in RDSon calculation). We choose standard value RCSENSE = 560 . OUT2:
Equation 40
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Design guidelines Equation 41
PM6680A
(Let's assume Tmax=75 C in RDSon calculation). We choose standard value RCSENSE = 560 .
5. Input capacitor
Maximum input capacitor RMS current is about 1.084 A. Then ICINRMS > 1.084 A We put two 10 F ceramic capacitors with Irms = 1.5 A.
6. Synchronous rectifier
OUT1: Shottky diode STPS1L40M OUT2: Shottky diode STPS1L40M
7. Integrator loop
(Refer to figure 14) OUT1: The ripple is smaller than 40 mV, then the virtual ESR network is required. CINT = 1.5 nF; Cfilt = 47 pF; RINT = 1.1 k OUT2: The ripple is smaller than 40 mV, then the virtual ESR network is required. CINT =1.5 nF; Cfilt =47 pF; RINT = 820
8. Output feedback divider
(Refer to figure 6) OUT1: R1 = 10 k; R2 = 27 k OUT2: R1 = 10 k; R2 = 10 k
9. Layout guidelines
The layout is very important in terms of efficiency, stability and noise of the system. It is possible to refer to the PM6680A demoboard for a complete layout example. For good PC board layout follows these guidelines:
Place on the top side all the power components (inductors, input and output capacitors, MOSFETs and diodes). Refer them to a power ground plan, PGND. If possible, reserve a layer to PGND plan. The PGND plan is the same for both the switching sections. AC current paths layout is very critical (seeFigure 41). The first priority is to minimize their length. Trace the LS MOSFET connection to PGND plan as short as possible. Place the synchronous diode D near the LS MOSFET. Connect the LS MOSFET drain to the switching node with a short trace. Place input capacitors near HS MOSFET drain. It is recommended to use the same input voltage plan for both the switching sections, in order to put together all input capacitors. Place all the sensitive analog signals (feedbacks, voltage reference, current sense paths) on the bottom side of the board or in an inner layer. Isolate them from the power top side with a signal ground layer, SGND. Connect the SGND and PGND plans only in one point (a multiple vias connection is preferable to a 0 ohm resistor connection) near the PGND device pin. Place the device on the top or on the bottom size and connect the exposed pad and the SGND pins to the SGND plan (see Figure 41).
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PM6680A
Phase HS CIN LS D
Place input capacitors together
L
Reduce the AC current paths
PGND plan (top layer)
COUT +
Low side gate trace (bottom layer)
CSENSE dedicated trace (bottom layer)
SGND2 PGND SGND2 LGATE5 LGATE1
Very close HGATE and PHASE traces (inner or bottom layers)
CSENSE5 CSENSE1 PHASE5 PHASE1 HGATE5 HGATE1 SGND1
SGND plan (inner layer)
Device (top layer) SGND connection to SGND plan
Multiple vias between SGND plan and PGND plan
Exposed pad connection to SGND
Signal traces
L LS
PM6680 HS
PGND
Figure 41. Current paths, ground connection and driver traces layout
Top layer PGND plan SGND plan
Bottom layer
Design guidelines
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Design guidelines
PM6680A
As general rule, make the high side and low side drivers traces wide and short. The high side driver is powered by the bootstrap circuit. It's very important to place capacitor CBOOT and diode DBOOT as near as possible to the HGATE pin (for example on the layer opposite to the device). Route HGATE and PHASE traces as near as possible in order to minimize the area between them. The Low side gate driver is powered by the 5 V linear regulator output. Placing PGND and LGATE pins near the low side MOSFETs reduces the length of the traces and the crosstalk noise between the two sections. The linear regulator output LDO5 is referred to SGND as long as the reference voltage Vref. Place their output filtering capacitors as near as possible to the device. Place input filtering capacitors near VCC and VIN pins. It would be better if the feedback networks connected to COMP, FB and OUT pins are "referred" to SGND in the same point as reference voltage Vref. To avoid capacitive coupling place these traces as far as possible from the gate drivers and phase (switching) paths. Place the current sense traces on the bottom side. Using It is recommended to use a dedicated connection between the switching node and the current limit resistor RCSENSE.

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PM6680A
Package mechanical data
10
Package mechanical data
In order to meet environmental requirements, ST offers these devices in ECOPACK(R) packages. These packages have a Lead-free second level interconnect. The category of second Level Interconnect is marked on the package and on the inner box label, in compliance with JEDEC Standard JESD97. The maximum ratings related to soldering conditions are also marked on the inner box label. ECOPACK is an ST trademark. ECOPACK specifications are available at: www.st.com.
Table 16.
VFQFPN 5x5 mechanical data (mm)
Dim A A1 A3 b D D2 E E2 e L ddd 0.30 4.85 0.18 4.85 Min 0.80 0 Typ 0.90 0.02 0.20 0.25 5.00 See exposed pad variations 5.00 See exposed pad variations 0.50 0.40 0.50 0.05
(1) (1)
Max 1.00 0.05
0.30 5.15
5.15
1. Dimensions D2 & E2 are not in accordance with JEDEC.
Table 17.
Exposed pad variations
D2 E2 Max 3.20 Min 2.90 Typ 3.10 Max 3.20
Min 2.90
Typ 3.10
Note:
1 2
VFQFPN stands for Thermally Enhanced Very thin Fine pitch Quad Flat Package No lead. Very thin: A=1.00 mm Max. Dimensions D2 & E2 are not in accordance with JEDEC.
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Package mechanical data
PM6680A
Figure 42. Package dimensions
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PM6680A
Revision history
11
Revision history
Table 18.
Date 12-Oct-2006 17-Dec-2007
Document revision history
Revision 1 2 Initial release. Added Section 5: Typical operating characteristics on page 12 and Section 9: Design guidelines on page 30 Changes
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PM6680A
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